In the design of operational amplifiers, it is important to provide a highly linear (i.e., low distortion), low noise amplifier capable of wide bandwidth operation. Bandwidth limitations, noise, and distortion can arise at any stage within the operational amplifier, but for present purposes the focus is upon the input stage. The typical input stage is a transconductor or transconductance circuit operable to convert an input voltage signal into an internal current signal more suitable for amplification by the output stage. Hence, the defining feature of the transconductance circuit is its voltage to current transfer function.
Prior Art FIG. 1 illustrates the prototypical input stage transconductor 10, i.e., a differential transistor pair. The transconductor 10 includes a pair of transistors Q1 and Q2 whose emitters are coupled to, a bias current source I.sub.DC that provides "tail" current for the transconductor 10. The differential voltage input pair V.sub.IN+ and V.sub.IN- drive the bases of the transistors Q1 and Q2, essentially steering the resulting differential current pair I.sub.OUT+ and I.sub.OUT- to a common ground reference 20. As will be appreciated, the voltage to current transfer function of the differential pair transconductor 10 is ideally a hyperbolic tangent (tanh) function.
While widely applicable and well suited for certain applications, the transconductor 10 suffers many shortcomings. When used within an amplifier having a capacitive feedback loop, as is often the case, the transconductor 10 is extremely limiting on the slew rate. (An amplifier's slew rate defines the maximum rate of change in voltage across the input and output terminals of the amplifier.) Specifically, the total current available to charge the feedback loop compensation capacitor C.sub.C is limited by the so-call "tail current" of the differential pair, i.e., the bias current I.sub.DC.
For the present analysis, it is fair to assume that the slew rate is equal to I.sub.DC /C.sub.C. Hence to improve the slew rate, one must decrease C.sub.C and/or increase I.sub.DC, both of which are undesirable for a variety of well known reasons. Additionally, the tanh transfer function of the differential pair transconductor 10 means that transconductor 10 is a non-linear, distortive circuit.
One common approach for addressing the slew rate limitations of the differential pair transconductor 10 of FIG. 1 is to use a class AB transconductance amplifier. Prior Art FIG. 2 illustrates one typical class AB amplifier 100 formed from a pair of differentially coupled diamond followers whose output emitters are coupled through a common load resistance R.sub.DGEN. Each diamond follower includes a pair of bias current sources I.sub.DC, and four transistors (one follower is made of transistors Q1-Q4, the other follower is made of transistors Q5-Q8).
The voltage to current transfer function of the class AB amplifier 100 without a common load resistance R.sub.DGEN (i.e., R.sub.DGEN =0) is ideally a hyperbolic sine (sinh) function. Prior Art FIG. 4 illustrates such an ideal transconductance of the class AB amplifier 100 (i.e., dIout/dVout) as a function of input voltage. As seen in FIG. 4, the ideal transconductance of the class AB amplifier 100 is non-linear at voltages close to zero, but fairly linear elsewhere. The transfer function of the class AB amplifier will vary for different values of R.sub.DGEN, but the non-linear characteristics are similar and related to the sinh function represented in FIG. 4.
In practice, the transconductance gain of the class AB amplifier 100 is set by the available bias current, the common load resistor R.sub.DGEN, and the nonlinear transconductance characteristics of the individual transistors. However, when R.sub.DGEN is large it dominates the nonlinear effects of the individual transistors, thereby improving the distortion characteristics of the class AB amplifier 100. Unfortunately, increasing R.sub.DGEN increases noise in the class AB amplifier 100 due to thermal noise of the resistor.
As mentioned above with reference to the differential pair transconductor 10 of FIG. 1, much of the non-linearity of transconductor 10 is due to the tanh nature of its transfer function. One well-known technique for linearizing differential pair transconductors is the so-called "multi-tanh technique." As will be appreciated, the key to the multi-tanh technique is the placement of multiple nonlinear tanh transconductors (i.e., differential pairs) along the input-voltage axis to achieve in combination a more linear transfer function.
Prior Art FIG. 3 illustrates a multi-tanh doublet 200 formed from two differential pairs Q1-Q2 and Q3-Q4 and two bias current sources I.sub.DC. Positive and negative offsets are introduced by forming each differential transistor pair with a gain imbalance. Specifically, a positive offset is introduced into the differential pair Q1-Q2 by forming transistor Q1 with a gain A that is greater than unity, and transistor Q2 with a gain of substantially unity. Likewise, a negative offset is introduced into the differential pair Q3-Q4 by forming transistor Q4 with a gain A that is greater than unity and transistor Q3 with a gain of substantially unity. Prior Art FIG. 5 illustrates the combined transconductance gain.
The multi-tanh transconductors do improve the distortion characteristics of an input stage, however the multi-tanh technique does not address the slew rate and other problems of the differential pair transconductor. Likewise, the class AB amplifier provides an improved slew rate, yet suffers from the nonlinearity about zero due to its sinh transfer function. What are needed are a variety of transconductance circuits that are highly linear with low noise, and having bandwidth characteristics not limited by slew rate.